Controller for a resonant converter

ABSTRACT

Consistent with an example embodiment, there is a controller for a resonant converter, wherein the controller is configured to operate the resonant converter in a high power mode of operation by adjusting a first control parameter to vary the output power and a low power mode of operation by adjusting a second control parameter to vary the output power. The controller is configured to set the value of the first control parameter when changing over between the high power mode of operation and the low power mode of operation such that the output power is substantially consistent during the changing over.

This application claims the priority under 35 U.S.C. §119 of Europeanpatent application no. 10252219.0, filed on Dec. 23, 2010, the contentsof which are incorporated by reference herein.

FIELD OF THE INVENTION

The present disclosure relates to the field of controllers for resonantconverters, and in particular, although not exclusively, to controllersthat can set control parameters for a changeover between a high powermode of operation and a low power mode of operation of the resonantconverter.

BACKGROUND OF THE INVENTION

WO 2009/098640 (NXP B.V.) discloses a method of operating a resonantpower converter in a low power mode of operation in which the switchingis controlled to allow for improved operation at low power levels. Themethod involves introducing an interruption to the part of the switchingcycle in which the low side switch of the resonant power converter wouldnormally be closed.

The listing or discussion of a prior-published document or anybackground in the specification should not necessarily be taken as anacknowledgement that the document or background is part of the state ofthe art or is common general knowledge.

SUMMARY OF THE INVENTION

According to a first aspect of the invention, there is provided acontroller for a resonant converter, wherein the controller isconfigured to operate the resonant converter in:

a high power mode of operation by adjusting a first control parameter tovary the output power; and

a low power mode of operation by adjusting a second control parameter tovary the output power; and

wherein the controller is configured to set the value of the firstcontrol parameter when changing between the high power mode of operationand the low power mode of operation such that the output power issubstantially consistent during the changeover.

Setting the value of the first control parameter when changing modes ofoperation in this way can reduce any discontinuity in the output powerat the changeover. For example, as a change in the second controlparameter is introduced for the low power mode of operation, the firstcontrol parameter is set so as to take the second control parameter intoaccount and maintain a substantially consistent output power.

The first control parameter may be an energy per cycle parameter. Thesecond control parameter may be a period time (Tper) parameter.

The controller may be configured to keep the energy per cycle levelconstant for the low power mode of operation. In this way, an efficientand economical low power mode of operation can be implemented with anoptimal/desired energy per cycle setting. It can be beneficial to varyonly one control at a time in order to maintain a simple linearrelationship between control input and power.

The controller may be configured to operate the resonant converter inthe low power mode of operation by adjusting the first control parameterand second control parameter to vary the output power. In one example,the output power can be set by adjusting both the first and secondcontrol parameters for power levels above a threshold level, such as aclamping voltage, and the output power can be set by adjusting only thesecond control parameter for power levels below a threshold level. Inthis way, the first control parameter can be brought down to adesired/optimum level during the low power mode of operation, and thenfixed at that level for even lower output power levels.

The controller may be configured to operate the resonant converter inthe low power mode of operation by:

adjusting only the first control parameter to vary the output power foroutput power levels above a threshold level, and

adjusting the second control parameter, and in some examples only thesecond control parameter, for output power levels below the thresholdlevel.

The controller may be configured to operate the resonant converter inthe low power mode of operation by:

adjusting both the first control parameter and second control parameterto vary the output power for output power levels above a thresholdlevel, and

adjusting the second control parameter, and in some examples only thesecond control parameter, for output power levels below the thresholdlevel.

The controller may be configured to multiply the first control parameterby a factor N when changing between the high power mode of operation andthe low power mode of operation. N may correspond to the ratio of theperiod time (Tper) to the period of energy conversion pulses (Tec) thatwill be employed in the low power mode of operation.

The period time (Tper) may be the minimum period time that can be usedin the low power mode of operation, as in some embodiments this is thevalue that will be used immediately after the changeover from the highpower mode of operation to the low power mode of operation.

In some examples, a low power mode of operation with an “energy dump”can be used, and this can affect the minimum value for the period time(Tper) that can be achieved and therefore can also affect the value ofN.

Further comprising an input representative of the number of energyconversion pulses (P) that are to be included in the period of energyconversion pulses. The number of energy conversion pulses (P) can affectthe period of energy conversion pulses (Tec). In this way, a user inputcan be provided to control the value of N that will be used as theperiod of energy conversion pulses (Tec) will vary in accordance withthe number of energy conversion pulses (P) that are used.

The controller may be configured to provide switch control signals tothe resonant converter in order to control the output power inaccordance with the control parameters of the high or low power mode ofoperation.

The controller may be configured to apply a minimum time intervalbetween successive power mode changes. In this way, unnecessaryrepetitive changeover between power modes can be reduced.

The controller may be configured to prevent a changeover between highpower mode and low power mode during one or more specific sub-states ofa mode timing sequence, which may be a high power mode sequence. Such achangeover can be from the high power mode to the low power mode, orfrom the low power mode to the high power mode. In this way, thelikelihood of introducing a discontinuity in the output power whenchanging mode of operation can be reduced.

The controller may be configured to operate the resonant converter in aburst mode of operation, wherein the converter is configured to preventthe switching frequency within the burst on time from corresponding toan audible frequency of the human ear. The controller may be configuredto prevent the frequency of operation in burst mode from dropping belowabout 20 kHz.

There may be provided a resonant converter comprising any controllerdisclosed herein.

There may be provided an integrated circuit comprising any controller,resonant converter or circuit disclosed herein.

According to a further aspect of the invention, there is provided amethod of operating a resonant converter, wherein the resonant converteris operable in:

a high power mode of operation by adjusting a first control parameter tovary the output power; and

a low power mode of operation by adjusting a second control parameter,or first control parameter, or first and second control parameter, tovary the output power; the method comprising:

setting the value of the first control parameter when changing betweenthe high power mode of operation and the low power mode of operationsuch that the output power is substantially consistent during thechangeover.

The low power mode of operation may involve adjusting one or more of:

the second control parameter,

the first control parameter, and

first and second control parameter,

for different output power levels.

At the changeover from the high power mode to the low power mode, thesecond control parameter (such as period time (Tper)) may be introduced,and require the first control parameter to be set so as to keep theoutput power substantially consistent during the changeover, even if thesecond control parameter is not adjusted for the output power levelsthat occur when the resonant converter first enters the low power mode.

There may be provided an integrated circuit comprising any controller,converter, circuit or apparatus disclosed herein.

There may be provided a computer program, which when run on a computer,causes the computer to configure any apparatus, including a controller,converter, circuit or apparatus disclosed herein or perform any methoddisclosed herein. The computer program may be a software implementation,and the computer may be considered as any appropriate hardware,including a digital signal processor, a microcontroller, and animplementation in read only memory (ROM), erasable programmable readonly memory (EPROM) or electronically erasable programmable read onlymemory (EEPROM), as non-limiting examples. The software may be anassembly program.

The computer program may be provided on a computer readable medium suchas a disc or a memory device, or may be embodied as a transient signal.Such a transient signal may be a network download, including an internetdownload.

BRIEF DESCRIPTION OF THE DRAWINGS

A description is now given, by way of example only, with reference tothe accompanying drawings, in which:

FIG. 1 illustrates schematically a series resonant converter;

FIG. 2 illustrates schematically an LLC converter;

FIG. 3 illustrates schematically an LCC converter;

FIG. 4 shows a simulation result of a known burst mode of operation foran LLC converter;

FIG. 5 illustrates a resonant converter and a controller according to anembodiment of the invention;

FIG. 6 illustrates graphically the operation of a resonant converterthat is controlled according to an embodiment of the invention;

FIG. 7 illustrates graphically the performance of a resonant converterin a low power mode of operation according to an embodiment of theinvention;

FIG. 8 illustrates the primary current and magnetizing current for a lowpower mode of operation according to an embodiment of the invention;

FIG. 9 illustrates schematically an implementation that can be used toadjust the energy per cycle according to an embodiment of the invention;

FIG. 10 illustrates graphically the operation of the resonant converterin a low power mode according to an embodiment of the invention;

FIG. 11 illustrates graphically the operation of the resonant converterin a low power mode according to another embodiment of the invention;

FIG. 12 illustrates graphically a function that regulates the operatingfrequency in a low power mode according to an embodiment of theinvention;

FIG. 13 illustrates schematically a circuit that can implement thefunctionality of FIG. 12;

FIG. 14 illustrates graphically the operation of a resonant converterthat is controlled according to an embodiment of the invention; and

FIG. 15 illustrates graphically the operation of a resonant converterthat is controlled according to another embodiment of the invention.

One or more embodiments of the invention relate to a resonant converterthat can operate in a high power mode of operation by adjusting a firstcontrol parameter (such as “energy per cycle”) to vary the output powerand a low power mode of operation by adjusting a second controlparameter (such as period time (Tper)) to vary the output power. Acontroller can be provided to set the value of the first controlparameter when changing between the high power mode of operation and thelow power mode of operation such that the output power is substantiallyconsistent during the changeover. The first control parameter can be setto take into account any fluctuation in power that would otherwiseoccur, for example by a change in, or introduction of the, secondcontrol parameter.

It is known to use resonant converter topologies for power convertersoperating at powers levels larger than approximately 100 Watt at fullload, due to their high efficiency and small volumes/high power density.

There are several known types of resonant converters, using either halfbridge or full bridge configurations. Also, the number of resonantcomponents can be different for different types of resonant converters.The series resonant converter forms the basis for other resonanttopologies, and a general circuit diagram of a series resonant converter100 is provided as FIG. 1.

The resonant components of the series resonant converter 100 of FIG. 1are a capacitor Cr 102 and an inductor Ls 104. As the magnetizinginductance of the transformer 106 is relatively large, it does notsignificantly influence the resonant cycle.

There is a known desire to have a low voltage across the high side andlow side switches 108, 110 when they are switched on in order to reduceturn on losses. This can be known as “soft switching”. Due to the largevalue of the magnetizing inductance of transformer 106 in the seriesresonant converter 100, the stored energy in the magnetizing inductanceis not sufficient to provide for soft switching. The current in theinductor Ls 104 is therefore necessary to obtain soft switching.

Multi resonant converters are derived from the basic series resonantconverter 100 shown in FIG. 1. The term ‘multi’ refers to the fact thatmore than two components contribute to the resonance. One type of multiresonant converter is the LLC converter 200 as shown in FIG. 2.

For the LLC converter of FIG. 2, the magnetizing inductance of thetransformer 206 also contributes to the resonance and is shown in FIG. 2as inductor Lm 212. Therefore, the resonant circuit (also known as aresonant tank) for the LLC converter of FIG. 2 consists of two inductors(Lm 212 and Ls 204) and one capacitor (Cr 202). This configurationallows the resonant converter to operate at a frequency that is belowthe resonant frequency of the series resonant converter 100 of FIG. 1 ina so called discontinuous mode. The magnetizing inductance Lm 212 allowsfor soft switching when the diodes 214, 216 at the secondary side of thetransformer 206 are not conducting.

Another type of multi resonant converter is an LCC converter 300 asshown in FIG. 3. A second resonant capacitor Cp 318 is connected inparallel with the secondary winding of the transformer 306 in the LCCconverter 300 of FIG. 3. The parallel capacitor Cp 318 can give a lowoutput voltage at high switching frequency, while the LLC convertergives a fixed output voltage at high switching frequency.

The LLC converter is most often used for regulation of output power whena fixed output voltage is necessary, and is typically controlled with a50% duty cycle at the half bridge node between the high side switch andlow side switch. The switching frequency can then be varied in order toregulate the output power. This method gives an acceptable efficiencyfor medium to large loads, however, for low loads, there is a drawbackof a relatively large circulating current, which results in a reducedefficiency at low load. It is known to improve this efficiency at lowloads by using a burst mode of operation, where the 50% duty cycle modeis applied during a burst on-time. After the burst on-time, both of thehigh side and low side switches (which are also known as half bridgeswitches) are turned off. A problem with known burst modes is associatedwith setting the proper frequency, because the frequency-power relationis steep and the tolerance of components affects the resonancefrequency, which in turn can lead to an inaccurately defined outputpower level. It is known in the art to set the frequency during a burstat a level that is higher than the normal operating frequency to accountfor the inaccurately defined output power level. A simulation result ofa known burst mode of operation for an LLC converter is shown as FIG. 4.

The simulation result of FIG. 4 shows two graphs. The top graphillustrates (i) the current through the magnetizing inductance of thetransformer, and is labelled Imagnetize 402; and (ii) the currentthrough the primary winding of the transformer, which is labelled Iprim404. The bottom graph shows the output current Iout 406.

It can be seen from FIG. 4 that, after the start of the burst, theoutput current Iout 406 is low, and therefore the steady state outputpower level is also low. This occurs after the time indicated as 320usec in FIG. 4. It can be seen that a relatively large power istransferred to the output during the initial cycles of the burst due tothe transient caused by the initial voltage at the resonant capacitor.Use of this burst mode gives an increased efficiency compared with acontinuous mode of operation, however the increase is limited to amaximum efficiency of approximately 75%.

WO2009/098640 discloses the setting of an “energy per cycle” level in alow power mode of operation in order to improve efficiency. In addition,circulating magnetizing current is prevented as much as possible bytemporarily storing magnetizing energy in the resonant capacitor incombination with an energy dump to the load, further improvingefficiency.

In many applications, the output of the converter is regulated to adesired value using a feedback loop that compares the desired value withthe sensed value, in order to generate an error signal in accordancewith the difference.

One or more embodiments disclosed herein can provide a resonant powerconverter with one or more of:

-   -   a high efficiency over load, which can include good efficiency        at very small load at an acceptable audio noise level;    -   stable control for output power, output voltage, output current        between no load and maximum load at nominal output voltage;

Embodiments disclosed herein can provide an advantageous way of changingover between different modes of operation of a resonant converter and/orsetting the energy level during a power conversion cycle. Such aresonant converter can be considered as improved over a completevariation of parameters.

FIG. 5 illustrates a resonant converter 502 and a controller 504according to an embodiment of the invention. The controller 504 providesswitch control signals to operate the switches of the resonant converter502 in accordance with one or more control parameters in order tooperate the resonant converter in either a high power mode of operationor a low power mode of operation. The controller 504 is configured toset the value of a control parameter associated with the low power modeof operation when changing between the high power mode of operation andthe low power mode of operation such that the output power issubstantially consistent during the changeover. This can prevent, orreduce the likelihood of, repetitive changeover between the modes ofoperation as the output power fluctuates at changeover, which can reduceaudio noise and also ripple voltage at the output. Examples of ways toimplement this functionality are described below.

It is known from WO2009098640 to operate the resonant converter in ahigh power mode where only the energy per cycle parameter can be variedin order to adjust the output power, and also in a low power mode whereboth energy per cycle and period time parameter can be varied in orderto adjust the output power. However, the period time makes a sudden jumpto a longer value when changing from high power mode to low power mode.Embodiments of the present invention can include the application of ascaling factor to set the converted energy per cycle for a low powermode of operation in order to compensate for this sudden jump. In thisway the output power in low power mode can be made approximately equalto, or overlap with, the output power in high power mode at thechangeover between modes of operation. In addition to this scalingfactor, the energy per cycle and period time parameter can be varied toregulate the output power in low power mode.

Embodiments disclosed herein can use a low power mode whereby the poweris regulated by adjusting (i) the energy per cycle; or (ii) theswitching frequency; or (iii) both.

The “energy per cycle” will now be described with reference to theseries resonant converter of FIG. 1, although it will be appreciatedthat the “energy per cycle” can also be defined for other types ofconverters. The energy per cycle can also be considered as the energyput into the resonant tank from the power supply and is related to thecharge transferred from the power supply to the resonant tank during theinterval that the resonant tank is connected to the supply. Thiscorresponds to a time when the high side switch 108 in FIG. 1 is closed,and causes a voltage difference deltaV across the resonant capacitor Cr102. As Q=Cr×deltaV, this value of deltaV corresponds to a certaincharge. The average current I drawn from the supply over a completecycle is then:I=Q/Tper=Q×Fswitch=Cr×deltaV×Fswitch;and the power taken from the supply is then:P=I×Vsupply=Cr×deltaV×Fswitch×Vsupply.As power is energy per time, the energy per cycle E:E=P×Tper=P/Fswitch=Cr×deltaV×Vsupply.

As the average voltage across Cr 102 cannot change in steady state,deltaV during the opposite switch conduction cycle will be identical butwith the opposite sign. According to the prior art, the energy per cycleis not set directly for high power mode with frequency control, but isrelated to the frequency of operation of the switches 108, 110 that isapplied. By using deltaV as a direct control criterion for driving theswitches (which may be referred to as capacitor voltage control) theenergy per cycle can be set directly. This method of capacitor voltagecontrol is therefore very suited to drive both low and high power modesaccording to embodiments of the invention, although is in no way theonly way of voltage control that can be used with embodiments of theinvention.

When reducing the output power, there is a desire to keep the efficiencyas high as possible. In some examples, it has been found that aparticularly good efficiency in low power mode is achieved by applyingan optimum, fixed, value for the energy per cycle setting and varyingthe switching frequency. Using such a value for the energy per cycle cangive a good compromise between RMS losses and core losses that issubstantially independent of the period time that is used. In light ofthe above, it can be considered advantageous in some examples toregulate the output power in low power mode of operation by adjustingthe switching frequency and maintaining the converted energy per cycleat a constant value.

In other examples it is considered advantageous to regulate the outputpower in low power mode of operation by adjusting one or both of theparameters: (i) the switching frequency and (ii) the converted energyper cycle. This can for example be advantageous when the energy percycle at the changeover point from high power mode to low power mode isat a value that is above the optimum efficiency setting. When the poweris regulated down in low power mode it can be beneficial to firstregulate the energy per cycle down to the optimum efficiency pointwhilst keeping the switching frequency fixed, and then keep the energyper cycle at the optimum efficiency level and reduce the switchingfrequency to reduce the output power further.

Embodiments of the invention disclosed herein can include a resonantconverter that can operate in one or more of:

-   -   a high power mode with power regulation by energy per cycle        control;    -   a low power mode with a fixed value for the energy per cycle,        whereby the output power is regulated by adapting the period        time; and    -   a low power mode whereby the output power is regulated by        adapting the period time and the energy per cycle. This low        power mode may be used in addition to the preceding low power        mode, for example to bring an energy per cycle value at the        changeover down to a preferred value, after which the low power        mode can be used with the fixed value for the energy per cycle.        whereby a changeover from the high power mode to a low power        mode occurs at a certain energy per cycle level. At this        changeover the energy per cycle value that was being used in the        high power mode is multiplied by a predetermined factor in order        to determine the value for the energy per cycle that will be        used at the start of the low power mode of operation. In this        way, any power step due to the step increase of the period time        can be compensated for. The predetermined factor is selected        such that any discontinuity in the output power when changing        from the high power mode of operation to the low power mode of        operation can be reduced when compared with the prior art.

In some examples any discontinuity when changing between modes ofoperation can be overcompensated for, for example by multiplying theenergy per cycle by a factor that overcompensates for the step increaseof the period time at changeover. In this way, the output power wouldthen be too large and the regulation loop further reduces the power.This overcompensation therefore fits within the goal of keeping theoutput power substantially consistent. This is also a way to build insome hysteresis in the changing between modes of operation of theresonant converter.

Embodiments of the invention are described in relation to varying a loadfrom a maximum value to a no/zero load value. The output of theconverter can be a voltage, current or power, depending on the actualapplication and load conditions.

FIG. 6 illustrates graphically a relationship between a control variableon the horizontal axis and the output power on the vertical axis for aresonant converter that is controlled according to an embodiment of theinvention. FIG. 6 illustrates the performance of the resonant converteras the control variable reduces from a maximum value towards zero. Inthis example an LLC resonant converter is used and the control variableis a control voltage (Vcontrol) that can adapt the output power (outputcurrent) between zero and full load over a voltage range betweenVcontrolmin 602 (which in this example is 1V) and Vcontrolmax 604. Itwill be appreciated that using a control variable in the voltage domain,as well as the range of values, are non-limiting examples of how theoutput power can be controlled.

The resonant converter that is represented by FIG. 6 operates in a lowpower mode of operation, which can be a low power mode of operation thatis disclosed in WO2009098640 and involve use of a fixed “energy percycle” value and a variable switching frequency. It will be appreciatedthat the “switching frequency” parameter relates to the inverse of thetime period of a complete switching cycle (Tper of FIG. 8).

As shown in FIG. 6 the resonant converter changes from the high powermode of operation to the low power mode of operation when the Vcontrolsignal drops below a threshold value V_HP-LP 606.

FIG. 6 also includes sketches of the shapes of the primary current 608and magnetizing current 610 of the transformer at different operatingpoints over a complete switching cycle.

In high power mode, a duty cycle of about 50% is often used for the ontime for both of the primary switches in the resonant converter,although embodiments of the invention are not limited in this regard. Asthe output power level is reduced in the high power mode of operation,the “converted energy per cycle” is reduced as can be seen from thedecreasing amplitude of the primary current 608. When the controlvariable (Vcontrol) is reduced such that it reaches the threshold levelV_HP-LP 606, the controller causes the resonant converter system tochange to the low power mode of operation. This is illustrated in FIG. 6as the primary current 608 does not oscillate for the complete switchingperiod, and the magnetization current 610 does not comprise a triangularwaveform when the primary current stops flowing, but instead startsringing.

It can be seen from FIG. 6 that the amplitude of the energy conversionpulses in the primary current 608 immediately increases after thechangeover from high power mode to low power mode. This is oneembodiment of how the output power can be kept substantially consistentat the changeover. This is because there is a larger time between energyconversion pulses in low power mode and therefore each energy conversionpulse needs to have a larger amplitude to maintain a similar timeaveraged output power.

It can also be seen from FIG. 6 that the amplitude of each energyconversion pulse in the primary current 608 is kept constant as thepower decreases in low power mode, and that the length of time betweensuccessive energy conversion pulses increases. This is indicative of theenergy per conversion cycle being kept constant and the switchingfrequency being reduced.

The low power mode represented by FIG. 6 will be described in moredetail with reference to FIG. 7, and is also described in W02009098640.In an example circuit (FIG. 1 and FIG.2 of W02009098640), the primarycircuit current 14 (I_(prim)), the voltage at the capacitor terminal 12(V_(cap)), the voltage at the half-brid node 10 (V_(hb)), output current16 (I_(out)), the high-side switch 19 (HSS) and the low-side switch 18(LSS) are depicted.

The low power mode of operation can be described in association with aso called ‘energy conversion interval’, which is represented by theintervals 24, 26 of FIG. 7 and is used to define the “energy per cycle”parameter disclosed herein. In FIG. 7, only one energy conversion pulsein the energy conversion interval is included in a complete sequence,but in general any integer number of energy conversion pulses ispossible. It is also possible to include a so called “energy dumpinterval” where at a predetermined point within the intervals 26, 28 thelow side switch is opened during a short period. During this shortperiod, magnetizing energy is converted to the load, giving theadditional advantage that during the rest of interval 26-34 and duringthe next 22 interval less core losses occur giving a higher efficiency.Under certain conditions it is even possible to skip the interval wherethe low side switch is closed again after the energy dump. In this casethe end of interval 34 is automatically reached.

For the low power mode of FIG. 6, two energy conversion pulses are usedwithin one complete switching cycle. This can be seen by the twocomplete oscillations in the primary current 608. This energy conversioninterval is the same as one complete period in high power mode. In FIG.6, one complete switching period is shown for the high power mode ofoperation. The output current (scaled to primary) is the differencebetween primary current 608 and magnetizing current 610.

It will be appreciated that the signals shown in FIG. 7 illustrate anon-limiting example of a low power mode of operation that can be usedwith embodiments of the present invention, and that other low powermodes, which may or may not use the same basic variables of energy percycle and Tper, can be used.

FIG. 8 illustrates the primary current 808 and magnetizing current 810for the low power mode of operation, and the primary current 808′ andmagnetizing current 810′ for the high power mode of operation, withequal levels set for the converted energy per cycle and magnetizingcurrent. That is, the amplitude of the primary current 808, 808′ is thesame in both the high power mode of operation and low power mode ofoperation. It will be appreciated that when the amplitude of the primarycurrents 808, 808′ are the same, the output power will be greater forthe high power mode of operation due to the pauses between energyconversion pulses in low power mode of operation.

The ratio between the average output current that is delivered to theload in high power mode and low power mode for the waveforms that areillustrated in FIG. 8 (that is, with the same energy per cycle) is:N=Tper/Tec  (1)

Where:

Tper is the period time in low power mode, and

Tec is the length of time that energy is converted.

During one example of a low power mode that can be used with embodimentsof the invention, it is desired to keep the converted energy per cycleat a substantially constant value in order to have good efficiency.Therefore, the time-averaged output power in low power mode is regulatedby adjusting the period time (Tper), while keeping the converted energyper cycle (and hence amplitude of the primary current) at a constantvalue.

In another example of a low power mode that can be used with embodimentsof the invention, it is desired to adjust the converted energy per cyclefrom a first value at changeover from high power mode to low power modeto a second value as the output power is reduced. The energy per cyclecan then be kept substantially constant at the second value for furtheroutput power reductions.

An embodiment of the invention relates to a manipulation of the energyper cycle value in for a low power mode of operation when changing froma high power mode of operation such that the time-averaged output powerchanges continuously during the changeover, and any discontinuities thatmay be introduced by the prior art are reduced.

In order to achieve the desired converted output power level at bothsides of the threshold from the high power mode to the low power mode(V_HP-LP 606 in FIG. 6), the step in power due to the ratio N iscompensated by multiplying the energy per cycle value for the low powermode with a factor N (according to the ratio of Tper/Tec) at the modechangeover. In this way, the energy per cycle is increased for low powermode by a factor that corresponds to the reduction in power that wouldhave occurred if the energy per cycle had been kept constant across thechangeover.

The frequency of operation (and hence Tper) just after entering the lowpower mode (that is close to the left-hand side of V_HP-LP 606 in FIG.6) is another parameter that affects the output power. It will beappreciated from the above discussion that Tper for the low power modemust be known in order to determine the correct value of N that shouldbe applied.

Looking to FIG. 7, and taking into account the resonant frequenciesduring the conversion interval while the secondary diodes are conducting(energy conversion intervals 24,26, Cr and Ls resonate) and the resonantfrequency during the intervals where the secondary diodes are notconducting (energy storage intervals 28-34 and energy restorage interval22 Cr and Ls+Lm resonate), a criterion can be set for the minimum ratioN close to the mode changeover. For this criterion, it is assumed that:

the total period of time covered by intervals 28-34, 22 is approximatelyequal to the resonant period time of Cr×(Ls+Lm), and

the total period of covered by intervals 24, 26 is approximately1.1×P×the resonant period time of Cr×Ls,

where P is the number of energy conversion intervals per complete lowpower cycle, and the constant factor of 1.1 is used to provide a marginfor the diode-off window during the energy conversion intervals. Thiscriterion can be considered as defining a minimum value for N as itinvolves consideration of the minimum period time at the modechangeover. In principle, any larger value for the minimum period timeis also possible, although may be less advantageous as larger peakcurrents would be necessary to achieve the same power level, which inturn causes extra RMS losses.

The interval 36 is used to regulate the period time (Tper) in the lowpower mode of operation, and therefore it needs to have a certainminimum value at the V_HP-LP border, which in some examples can takeinto account tolerances and design variations of Ls and Lm. Examplevalues for N for different values of P, including a maximum ratio Lm/Lsof 7 and that take into account the above mentioned considerations, aregiven in Table 1 below:

TABLE 1 Energy pulses/cycle (P) N 1 3.34 2 2.17 3 1.78

In order to further reduce the output power level when the resonantconverter enters the low power mode of operation, the period time (Tper)is increased. This is illustrated by the example signals for the primarycurrent 608 in the low power mode of operation. In principle the periodtime can be increased to an infinite value in this way in order to bringthe output power down towards zero.

FIG. 9 illustrates schematically an implementation that can be used toadjust the energy per cycle according to an embodiment of the invention.The “energy per cycle” is one example of a parameter that is controlledin order to adjust the output power.

The circuit of FIG. 9 receives:

a Vcontrol input signal 912 representative of the desired output power;

an HP input signal 906 representative of whether the resonant convertershould operate in a high power or low power mode of operation; and

onepulse and threepulse input signals 902, 904 representative of howmany energy conversion cycles are to be included in a switching cycle(Tper in FIG. 8) when operating in a low power mode of operation.

By processing the above input signals, the circuit of FIG. 9 provides anoutput signal at node Vdm 910 that is representative of the required“energy per cycle” value in order to operate according to an embodimentof the invention. The output signal at node Vdm 910 represents of theresult of a multiplication factor N (as determined by onepulse andthreepulse input signals 902, 904) being applied to the control signal(Vcontrol input signal 912) if a low power mode of operation is beingemployed (as determined by the HP input signal 906). If a low power modeof operation is not being employed, then a multiplication factor N of 1is used.

The Vcontrol signal 912 (which is an example of a control variable inthe voltage domain) is provided to a voltage to current converter, andthe resultant current is provided as an input to the current mirror 922.The voltage to current converter comprises an amplifier 924 thatreceives the Vcountrol signal 912. The output of the amplifier 924 isconnected to the gate of a FET 925. The conduction channel of the FET isconnected in series with a resistor 926 and a DC offset voltage source928 between ground and the input to the current mirror 922. It will beappreciated that any other type of voltage to current converter could beused.

The magnitude of the current that is provided to the input of thecurrent mirror 922, using the well known equation I=V/R, depends on:

the magnitude of the resistor 926, which in this example is R; and

the voltage across the resistor 926. In this example, the voltage acrossthe resistor 926 is the value of the control signal (Vcontrol 912) minusthe value of the DC offset voltage source 928. The DC offset voltagesource 928 of FIG. 9 has a value of 1V, which corresponds to the valueof Vcountrolmin 602 in FIG. 6.

The current mirror 922 provides an output current that is equal to theinput current; the current mirror provides a 1:1 ratio between its inputand output. The output current is provided to four resistors 914, 916,918, 920 in series. The Vdm output voltage value 910 can then be tappedoff from a position in the chain of resistors 914, 916, 918, 920 toprovide the required multiplication factor in accordance with the modeof operation of the resonant converter.

The resistance values of the four resistors 914, 916, 918, 920 areselected to provide the desired values for N. In this example, thevalues of N identified in Table 1 above are used and:

resistor 920 has a value R, which is the same as the resistance value ofthe resistor 926 at the input side of the current mirror 922 in order toprovide N=1;

resistor 918 has a value of 0.78×R so that the sum of the resistances ofresistors 920 and 918 equals 1.78×R, and N=1.78;

resistor 916 has a value of 0.39×R so that the sum of the resistances ofresistors 920, 918 and 916 equals 2.17×R, and N=2.17;

resistor 914 has a value of 1.16×R so that the sum of the resistances ofresistors 920, 918, 916 and 914 equals 3.34×R, and N=3.34.

The onepulse 902, threepulse 904 and HP 906 input signals are used tocontrol associated switches in order to connect the Vdm output node 910to one of the junctions between the four resistors 914, 916, 918, 920.In this way an appropriate multiplication factor can effectively beapplied to the Vcontrol signal 912 when the resonant converter isoperating in a low power mode of operation such that a more smoothhandover between the different modes of operation can be achieved. This,in turn, can reduce the likelihood of the resonant converter repeatedlyswitching back and forth between different modes of operation whenchanging mode.

The circuit of FIG. 9 receives an HP input signal 906 that isrepresentative of whether the system should operate in high power modeor low power mode. When the HP input signal is representative of a highpower mode of operation, the HP switch in FIG. 9 is in the lowerposition and the voltage across the single resistance 920 is tapped offas the Vdm output node 910. As the resistance value of resistor 920 isequal to the resistor 926 at the input side of the current mirror, thevoltage that is tapped of to the Vdm node 910 is equal to the voltageacross the resistor 926. As discussed above, the voltage that is droppedacross the resistor 926 at the input side is equal to Vcontrol minus theDC offset.

When the HP input signal is representative of a low power mode ofoperation, the HP switch in FIG. 9 is in the upper position, and thevoltage that is provided to the Vdm output can be:

the voltage across the sum of resistors 920 and 918 in order to apply amultiplication factor of 1.78. The multiplication factor of 1.78 isapplied when the threepulse signal 904 is active;

the voltage across the sum of resistors 920, 918 and 916 in order toapply a multiplication factor of 2.17. The multiplication factor of 2.17is applied when the onepulse and threepulse signals 902, 904 areinactive;

the voltage across the sum of resistors 920, 918, 916 and 914 in orderto apply a multiplication factor of 3.34. The multiplication factor of3.34 is applied when the onepulse signal 902 is active and thethreepulse signal 904 is inactive.

The circuit of FIG. 9 can operate with one, two or three energyconversion intervals, although any number of intervals can be used inother embodiments. FIG. 6 illustrates operation with two energyconversion cycles in low power mode, and FIG. 7 illustrates operationwith one energy conversion cycle.

The amplifier 924 also receives two threshold signals: Vclamp1 andVclamp2, which are used to limit the values of Vcontrol 912 that areconverted to the current that is passed through resistor 926. Vclamp1defines a lower limit of Vcontrol 912 and Vclamp2 defines an upperlimit. As illustrated in FIG. 9, the energy per cycle will be limited tothe level as set by vclamp1 if vcontrol is lower than vclamp(Highest(Vcontrol,Vclamp1)). This feature allows a reduction in theenergy per cycle level in low power mode until Vcontrol reaches Vclamp1,below which Vcontrol is held at vclamp1, and the energy per cycle iskept fixed. Also, the level of Vclamp2 is used to limit the energy percycle to a maximum level in low power mode (Lowest(Vcontrol, Vclamp2).The values for Vclamp1 and Vclamp2 can be set in accordance with:

the state of the HP input signal 906, which defines whether the resonantconverter is operating in a high or low power mode of operation;

the state of the onepulse and threepulse input signals 902, 904, whichdefine how many energy conversion pulses are used per cycle in the lowpower mode of operation; and

a “dump” parameter.

The “dump” parameter is a mode selection input for the logic part of theconverter that is used to select if an “energy dump” interval isincluded or not, and is known from prior art document WO2009098640. Forexample FIG. 7 of WO2009098640 illustrates an energy dump interval 150.Introducing an energy dump interval gives a slight change in the optimumenergy per cycle for getting maximum converter efficiency. This alsoholds for the number of energy conversion pulses selected. With energydump, the optimum efficiency can occur at a slightly lower energy percycle compared to a “no dump” implementation. Therefore Vclamp1 can bechosen slightly lower if ‘dump’ is selected, as the level of Vclamp1sets the lower limit for Vcontrol down to where the energy per cycle isreduced when Vcontrol is lowered.

The values for Vclamp1 and Vclamp2 are set so as to provide a high powermode of operation whereby the “energy per cycle” parameter is adjustedin order to change the output power, and a low power mode of operationwhereby a switching frequency and/or the energy per cycle parameter isadjusted to change the power. The energy per cycle parameter is keptconstant during the low power mode of operation for values of Vcontrolless than Vclamp1. Such operation is described below.

When the resonant converter is in the high power mode of operation (HP906 is active) Vclamp1 is set as a low value and Vclamp 2 is set as ahigh value that is outside the range of Vcontrol (912). In this way, thevalue for Vcontrol 912 is always between the two clamp values and iscopied to the source of the MOST follower 925. Therefore, in the highpower mode of operation, the value at the Vdm output node 910 isproportional to Vcontrol−Vref and the converted energy per cycle isvaried in order to adjust the output power of the resonant converter.

In one example, when the resonant converter is in the low power mode ofoperation (HP 906 is inactive), both Vclamp1 and Vclamp2 are set as avalue that corresponds to the voltage at which the resonant converterchanges from the high power mode of operation to the low power mode ofoperation. This changeover point is described above with reference toV_HP-LP 606 in FIG. 6. In this way, the desired fixed energy per cyclevalue is applied at the HP-LP changeover point and is then maintainedfor lower powers. It will be appreciated that in order to reduce theoutput power further (and maintain a constant value for the energy percycle), reducing the frequency of operation is one way of furtherreducing the output power. FIG. 10 illustrates graphically the operationof the resonant converter in a low power mode whereby both Vclamp1 andVclamp2 are set as a value that corresponds to V_HP-LP. It can be seenthat the value for Vdm, and hence the value for the energy per cycleparameter is fixed and independent of the output power that is required.In this way the values for Vclamp1 and Vclamp2 define the energy percycle in low power mode, and also when Vcontrol is greater than V_HP-LPif Vclamp1 and Vclamp2 are still applied.

FIG. 11 shows an alternative example whereby Vclamp1 is set at a valuebelow Vclamp2, and the value for Vclamp2 generally corresponds withV_HP-LP. In this way, the energy per cycle can be reduced when theresonant converter is operating in the low power mode of operation onlywhen the value for Vcontrol is greater than Vclamp1. When Vcontrol dropsbelow Vclamp1, the energy per cycle is fixed and the switching frequencymust be changed in order to reduce the output power further. Therefore,a low power mode can be entered that initially involves reducing theenergy per cycle, and then reducing the frequency and keeping the energyper cycle fixed.

This embodiment can be used when at the changeover point from high powermode to low power mode, the energy per cycle is larger than the optimumenergy per cycle. Therefore, this embodiment can be advantageous toreduce the energy per cycle during the low power mode until it reachesan optimum value, at which point it can be fixed. In such an example, itis possible to keep the period time fixed during the interval where theenergy per cycle is being reduced, although embodiments that vary boththe energy per cycle and period time (Tper) within this interval canalso be provided.

In another embodiment, Vclamp2 can be set as a value that is higher thanV_HP-LP. It is recalled that V_HP-LP is the changeover from high powermode to low power mode, and is not necessarily the same as thechangeover point from low power mode to high power mode. In thisexample, increasing Vcontrol above V_HP-LP will cause Vdm to beincreased whilst still operating in low power mode up until the maximumlevel is reached as defined Vclamp2. This embodiment can be advantageouswhen at the changeover point from high power mode to low power mode, theenergy per cycle is smaller than the optimum energy per cycle. Theadvantage lies in the fact that the energy per cycle can be increasedbefore the resonant converter changes back to a high power mode ofoperation. In this example it is possible to keep the period time fixedduring the interval where the energy per cycle is increased, althoughembodiments that vary both the energy per cycle and period time (Tper)within this interval can also be provided.

FIG. 12 illustrates graphically a function that regulates the operatingfrequency in low power mode. The functionality of FIG. 12 can beimplemented by the circuit of FIG. 13 as described below. The verticalaxis of the graph is the voltage at capacitor 1302 in FIG. 13, and thehorizontal axis is time. FIG. 12 shows how the current changes (by thedv/dt giving the slope of the voltage at the capacitor 1302) for animplementation that uses a single energy conversion pulse 1202, twoenergy conversion pulses 1204 and three energy conversion pulses 1206.In addition, the energy conversion period (Tec) of the single energyconversion pulse 1202 example is identified with reference 1208, and thetotal time period of the cycle (Tper) with a single pulse is identifiedwith the reference 1210. The difference between Tper 1210 and Tec 1208is referred to as the discharge time Tdis, and is identified in FIG. 12with reference 1212

It will be appreciated from the illustration of a low power waveform inFIG. 8, that the length of the interval Tec (during which energyconversion pulses are generated) is dependent on (i) the number ofenergy conversion pulses (P); and also (ii) the product of thecapacitance value of the resonant Cr and the inductance value of theresonant inductor Ls. The capacitance value of the resonant Cr and theinductance value of the resonant inductor Ls affect the frequency withwhich the resonant tank oscillates and therefore the duration of theenergy conversion pulses that are shown in FIG. 8.

The value of N (Tper/Tec) is dependent on (i) the number of energyconversion pulses (P); and (ii) the ratio of the inductance valuesLs/Lm. The value of N is not dependent on the capacitance value of theresonant capacitor Cr. It is necessary to make the discharge intervalTdis 1212 directly dependent on Tec in order for the desired value of Nat V_HP-LP to be maintained, even if the value of Tec is changed. If theresonant capacitor Cr is changed for various applications, then Tecchanges as it is directly the copied timing of the switches. Using FIG.7 for further explanation of one example: if Cr doubles, then theinterval 24+26 of FIG. 7 (which equates to Tec) is increased by 41%, andthe interval 28 to 34 (which equates to Tdis) also increases by 41% asit scales with Cr in the same way as Tec. This relationship requiresTdis to be directly proportional to Tec, which is automatically achievedby the circuit of FIG. 13 as the capacitor C 1302 is charged for a 41%longer time to a 41% higher voltage, and therefore so discharging alsotakes 41% longer. Furthermore, it is also necessary to make the intervalTdis 1212 a function of the control parameter in order to properlyregulate the power.

Although the value for Tec 1208 could be set externally by a user, thiswould require extra pins on an integrated circuit (IC) that houses theresonant converter and/or would require programming for setting therequired level. However, embodiments disclosed herein seek to integratethe functionality illustrated by FIG. 12 on-chip, and therefore it isbeneficial to avoid the use of additional pins and external programmingin such embodiments. Embodiments disclosed herein can provide on-chipadaptive setting of the Tdis parameter as a function of both Tec (inorder to keep N constant at the V_HP-LP border) and as a function ofVcontrol using Vclamp_a and Vclamp_b (in order to linearly increase Tdiswith Vcontrol when Vcontrol is lower than Vclamp_a). This enables Nto becorrectly set according to Table 1 when Vcontrol is equal to Vclamp_aand Vclamp_b.

As described above in relation to FIG. 10, both Vclamp_a and Vclamp_bare set equal to V_HP-LP when a fixed energy per cycle level is used andthe frequency is reduced in order to reduce the output power in lowpower mode. This enables N to be correctly set according to Table 1 whenVcontrol is equal to V_HP-LP.

Alternatively, if Vclamp_a and Vclamp_b are both set to a value that isbelow V_HP-LP, then this still gives the same N at the V_HP-LP border(both charging and discharging current for C1302 are reduced with thesame factor by reducing both Vclamp_a and Vclamp_b in the same way) butVcontrol is only influenced when Vcontrol is less than Vclamp_a (whichequals Vclamp_b), and this enables the functionality of reducingVcontrol by first keeping the frequency fixed and reducing the energyper cycle, and then keeping the energy per cycle fixed and furtherreducing the frequency. This reduction in frequency can be achieved byincreasing Tdis. In relation to the example of FIG. 9, thisfunctionality can be achieved by setting Vclamp1 of FIG. 9 to the valueof Vclamp_a and Vclamp_b. This gives the changeover from energy percycle reduction to frequency reduction atVcontrol=Vclamp_a=Vclamp_b=Vclamp1.

FIG. 12 illustrates that the slope of the increasing current isdifferent for each of the one pulse, two pulse and three pulse waveforms1202, 1204, 1206 such that the maximum current at the end of Tec 1208 isthe same for each of the one pulse, two pulse and three pulse waveforms1202, 1204, 1206. As will be described with reference to FIG. 13, theincreasing current represents a period of time during which a capacitoris being charged and the different slopes are enabled by introducingdifferent resistance values into the capacitor charging circuit.

During the discharge interval Tdis 1212, the capacitor is dischargedwith a current that depends on the control parameter (Vcontrol), butdoes not depend on the number of energy conversion pulses P. That is,the slope of the current during the Tdis interval is independent of thelength of the capacitor charging interval Tec 1208.

It can be advantageous for the slope of the charging current to be setat a value that is inversely proportional to P because it reduces themaximum voltage window of the capacitor 1302 due to variations in thevalues of Cr, Ls and P. In the example of FIG. 12, the slope of thecharging voltage for each waveform is:dv/dt=k1×(Vclamp_(—) b−Vref)/(P×R)]where R is the resistance value of each of the resistors in the chargingcircuit of FIG. 13. By using resistors with equal resistance values, theslope of the charging current can be set at a value that is inverselyproportional to P. In this way, the dependence on P of the factor N isautomatically realized. An additional advantage is that the maximumvoltage at the capacitor is almost constant for different P values.

The end of the cycle (Tper 1210) is defined by the current reaching thesame value as at the start of Tec 1208.

The functionality illustrated by FIG. 12 can be considered as adaptivebehaviour as the component values of the resonant tank affect the Tec1208 interval, which in turn affects the Tper 1210 interval, andtherefore also affects the output power. Therefore, the values of thecomponents in the resonant tank can be adjusted/adapted in order tochange the output power while maintaining the proper Tdis value. It willbe appreciated that in embodiments where the ratio Tper/Tec (N) is keptconstant, the Tper interval 1210 will also be adjusted as Tec interval1208 is changed.

If the desired output power increases whilst the converter is in the lowpower mode of operation, and the control parameter is allowed to risesufficiently, there will be a lower limit of the minimum period timethat can be implemented as the interval 36 in FIG. 7 becomes zero or aminimal value. At this point the converter must change to the high powermode of operation in order for the output power to be increased further.

In some embodiments, a low power mode of operation with energy dump canbe used as identified above and explained in WO02009098640 at page 25,line 32 to page 26, line 12. The result of this mode of operation isthat the intervals 28,30,32,34 identified in FIG. 7 are skipped andinterval 36 is almost immediately reached after the end of Tec. That is,the interval shown with reference deltaT 812 in FIG. 8 can be skipped.

The effect of this low power mode of operation is that the power can nowbe increased much further in low power mode. Therefore the levelVclamp_a can be set slightly higher than V_HP-LP, while Vclamp_b can beset equal to V_HP-LP. This prevents the frequency from being furtherincreased when Vcontrol gets above Vcontrol_a, but keeps the required Naccording to Table 1 at Vcontrol=V_HP-LP. This gives the criterion tochange back from low power mode to high power mode to occur atapproximately the same Vcontrol when this CCM energy dump is used.

FIG. 13 shows a functional schematic diagram for implementing thefunctionality of FIG. 12.

An integrator capacitor 1302 is used for realizing theintegration/timing illustrated by FIG. 12. It will be appreciated thatthis capacitor solution is only one specific embodiment, and that anyanalogue or digital circuit or computer algorithm that can perform anintegration or counting action in the time domain during the Tec andTdis (Tper-Tec) intervals with predefined factor in the integrationaccording to FIG. 12 can be used.

During the energy conversion interval Tec 1208, the capacitor 1302 ischarged by a current 2.34×(Vclamp_b−Vref)/(P×R), where P represents thenumber of pulses in the energy conversion period Tec, and R representsthe resistance value of each of the resistors 1304, 1306, 1308 describedbelow.

The circuit of FIG. 13 includes two different Vclamp levels (Vclamp_aand Vclamp_b), the significance of which is described below. At thisstage in the description of FIG. 13, the values of Vclamp_a and Vclamp_bcan be considered as sufficiently similar such that they can be referredto as having the same value of Vclamp.

The circuit of FIG. 13 includes three resistors 1304, 1306, 1308 ofequal value in series. Two input signals onepulse and twopulse are usedto control associated switches in order to selectively include or bypassthe resistors 1304, 1306, 1308 as part of a voltage to current converter1310. In this example, the voltage to current converter 1310 is used toconvert the Vclamp_b voltage to a current that is inversely proportionalto resistance of those resistors 1304, 1306, 1308 in the resistor chainthat are in use.

The current that is provided by the voltage to current converter 1310 isused as an input to a current mirror 1312 with a mirror ratio 2.34:1.The value of 2.34 is used to define a charge current that is 2.34 timesas large as the discharge current in order to give the voltage at 1302the shape according to FIG. 12 and N the correct value according toTable 1 when the lowest of Vcontrol and Vclamp_a equals Vclamp_b(lowest(Vcontrol,Vclamp_a)=Vclamp_b). The output of the current mirror1312 is used to charge the integrator capacitor 1302 during the energyconversion period Tec, and the output of the current mirror 1312 isdisconnected from the integrator capacitor 1302 after expiry of theenergy conversion period Tec using switch 1314. The charging current(Ich) can be defined as:Ich=2.34×(Vclamp_(—) b−Vref)/(P×R)

During the discharge interval Tdis (Tper−Tec), a second switch 1316 isclosed to provide a discharge path for the integrator capacitor 1302 tothe output of a 1:1 current mirror 1326. The capacitor is dischargedwith a current lowest(Vcontrol,Vclamp_a)−Vref]/R.

The left-hand side of the circuit of FIG. 13 includes a voltage tocurrent converter 1324 that receives Vcontrol and a lower clamping limit(Vclamp_a) as input signals. Connected in series with the output of thevoltage to current converter 1324 is a reference voltage source Vref1318. The output of the voltage to current converter 1324 is connectedto the input of a first 1:1 current mirror 1322, and the output of thefirst 1:1 current mirror 1322 is connected to the input of the 1:1current mirror 1326 that is identified above. These components 1318,1324, 1322 can be considered as defining the discharge current of thecapacitor.

The output of the circuit is the voltage across the capacitor 1302 asfunction of time, which is illustrated by FIG. 12, and can be used tostart the next energy conversion period Tec when the voltage acrosscapacitor 1302 reaches the same value as at the beginning of the Tecinterval.

As shown in FIG. 13, an extra DC voltage signal Vbias is subtracted fromthe offset DC voltage source Vref 1318 in order to provide an additionalbias current Vbias/R in the current mirrors. The additional bias currentcan be required in some embodiments to keep the current mirrors biasedwith a minimum current when Tper becomes large, thereby providing a longdelay between the energy conversion intervals Tec of successive cycles.Using this minimum current can reduce or prevent unwanted parasitic andoscillating behaviour. However, the additional discharge current candisturb the process of discharging the capacitor 1320. Therefore thesame bias current is separately added to the discharge current of thecapacitor 1302 by a further voltage to current converter 1320 that isused to convert the bias voltage level to a current level that isconsistent with the additional current introduced by the DC offsetvoltage source Vref 1318.

Further details of the use of two different Vclamp values (Vclamp_a andVclamp_b) will now be described.

When Vcontrol equals the value of Vref 1318+Vbias, the current at theoutput of the first current mirror 1322 will equal the current at theoutput of the voltage to current converter 1320 such that the currentscancel each other at the capacitor 1302. This in turn causes the periodtime in low power mode to be made very large and tend towards infinitywhilst maintaining a bias current for the current mirrors. By settingthe voltage level Vclamp_b of the second voltage to current converter1310 equal to the changeover point from high power mode to low powermode (V_HP-LP of FIG. 6), the desired ratio of Tper/Tec (N) is obtainedat the mode changeover.

The voltage clamp level Vclamp_a of the first voltage to currentconverter 1324 can be set at a value that is slightly higher thanVclamp_b (and therefore also slightly higher than V_HP-LP) such that theoutput power can be increased further by increasing Vcontrol (thecontrol variable) above the value V_HP-LP. This enables a furtherincrease in the frequency when the output power is being increased inlow power mode, and a reduction of N (Tper/Tec) as function of Vcontrol.This increase in the frequency can be made until the interval 36 shownin FIG. 7 becomes zero, a predefined minimal value, or until Vclamp_a ofthe first voltage to current converter 1324 is reached. These are allexamples of criteria for changing over to the high power mode ofoperation.

An embodiment for realizing the changeover to the high power mode ofoperation can utilise a regulation loop that increases the controlvariable during the low power mode of operation. There will come a pointat which the power cannot be further increased whilst remaining in thelow power mode of operation, and when vcontrol reaches the associatedthreshold value for the control variable the changeover to high powermode takes place. In this way hysteresis can be created between thechangeover point from high power to low power mode, and vice versa. Thisis shown in FIG. 14. It can also be possible to switch back to highpower mode when Vcontrol is larger than the threshold or in combinationwith a timer criterion. The effect is that the system can allow a lackof power to be delivered for a certain time, causing the regulatedoutput to drop slightly.

FIG. 14 includes all of the features of FIG. 6 in relation to areduction in the output power, and also illustrates the changeover fromthe low power mode of operation to the high power mode of operation asthe output power increases. The changeover from the low power mode ofoperation to the high power mode of operation is shown as V_LP-HP 1402in FIG. 14.

The hysteresis of FIG. 14 can be achieved by setting Vclamp_a 1402 ofthe first voltage to current converter 1324 of FIG. 13 to a level thatis above the V_HP-LP 1404 threshold for Vcontrol, while Vclamp_b for thesecond voltage to current converter 1310 of FIG. 13 is set at theV_HP-LP 1404 threshold for Vcontrol in order to get the optimum Naccording to Table 1 at the high power mode to low power mode changeoverpoint, while the frequency can still be increased when Vcontrolincreases above the V_HP-LP 1404 changeover point, thereby entering thehysteresis region.

The bold dotted line of FIG. 14, which is shown with reference numbers1407, 1408, 1410, 1412, shows the output power that can be produced foran increasing value of Vcontrol according to an embodiment of theinvention.

In some examples the resonant converter may not immediately change froma low power mode to a high power mode when the Vcontrol signal reachesthe V_LP-HP 1402 level. The same can be true for the changeover fromhigh power mode to low power mode. Two example reasons for the delay inchanging mode of operation are:

-   -   1. use of a timer that ensures that a minimum period of time is        spent in a mode of operation before changing again. Such a timer        can be started as soon as a mode of operation is entered and can        be used to prevent a further change of mode until the timer has        expired. This can be useful in allowing the regulation loop to        settle and for any overshoots to be dealt with.    -   2. use of a filter to process the Vcontrol signal so as to        reduce the effects of any ripple that may be present in the        control signal. Such a filter can inherently introduce a delay        between the instantaneous value of the control signal crossing a        threshold and the filtered signal reaching the same threshold.

An example of such a delay in changing from a low power mode ofoperation to a high power mode of operation will now be described withreference to FIG. 14. It will be appreciated that although the actualchangeover is shown as occurring at voltage 1406, the changeover is notdependent on the voltage at 1406, but instead is representative of aninstant in time when Vcontrol happens to be at the level of 1406. Adependence on time cannot be directly represented by FIG. 14.

As the control voltage increases whilst the resonant converter is in thelow power mode of operation, the output power gradually increases andfollows the line 1407 in FIG. 14. When Vcontrol reaches the Vclamp_a1402 value, any further increases in Vcontrol are not considered (suchfunctionality can be implemented by the circuit of FIG. 13 for example)and, as long as the resonant converter is in low power mode ofoperation, the output power cannot continue to increase. This isrepresented in FIG. 14 by the horizontal line 1408, which represents aconstant value for the output power.

The actual changeover to high power mode of operation occurs at 1406 inFIG. 14, and example timing issues are described above to explain whythis may not occur immediately after Vcontrol reaches V_LP-HP. When thehigh power mode of operation is entered, the output power jumps to avalue that would have occurred if the time delay had not been incurred,and this is represented by the vertical line 1410 in FIG. 14. The outputpower then continues to increase as Vcontrol increases as shown withreference 1412 in FIG. 14.

It will be appreciated that the embodiment of FIG. 13 is onenon-limiting example implementation of how to achieve the desiredfunctionality as shown in FIG. 12. Any other way of implementing thefunctionality described in relation to FIGS. 12 and/or 14 can beprovided. For example, circuitry operating in the current domain insteadof voltage domain can be used, and the Vcontrol signal can be replacedwith a current signal, switched resistors can be replaced with switchedcurrents, etc. In addition, the slope/ramp during Tec 1208 in FIG. 12could be made constant, while making the slope/ramp during Tdis 1212dependent on P in order to realize the proper N, digital implementationof the function, including counters.

When changing between low power mode and high power mode, or vice versa,it can be advantageous to allow only a changeover during certainsubintervals of the conversion cycle to reduce or prevent instantdisturbances of the resonant tank. For example, it can be advantageousfor the changeover from low power mode to high power mode to take placeduring state 36 in FIG. 7. It can also be advantageous for thechangeover from high power mode to low power mode to take place at theend of a complete cycle.

Also, it can be advantageous for the first cycle following a changeoverbetween low power mode and high power mode, or vice versa, to start at awell defined state. For example, at changeover from high power mode tolow power mode the first low power mode cycle should preferably startwith state 28 in FIG. 7 in order to get a natural changeover. Suchchangeover conditions can be considered as features of one or moreembodiments of the invention.

When a potential change between low power mode and high power mode, orvice versa, is identified it can be advantageous to prevent a changeoverif the system was not operating for at least a certain minimum time inthe current mode. A controller associated with the resonant convertercan apply a minimum time interval between successive power mode changes.This is an advantage as it allows the regulation loop to settle to thenew situation. Without this feature there is a risk that any transienteffects that may occur will trigger the opposite mode of operation. Thisfunctionality can be implemented in software associated with thecontroller, or with hardware.

When a potential change between low power mode and high power mode, orvice versa, is identified it can be advantageous to prevent a changeoverbetween high power mode and low power mode during one or more specificsub-state of the mode timing sequence, such as the high power modesequence. Examples of such specific sub-states are all intervals except24 and 26 shown in FIG. 7. Such sub-states can be indicative of aninstant in time when changing the mode of operation would generate adiscontinuity in the operation of the converter, and correspondingsub-states for different types of resonant converters and differentmodes of operation will be known to the person skilled in the art. Thisfunctionality can be implemented in software associated with thecontroller, or with hardware.

Burst Mode Operation

One or more embodiments disclosed herein can implement a burst mode ofoperation. There are several ways to implement burst mode in a switchedmode power converter, which are prior art, for example starting theburst when the regulated output voltage of the converter becomes lowerthan desired and stopping the burst when it becomes higher than desired.Another prior art solution is to start the burst when the regulatedoutput voltage of the converter becomes lower than desired and stop theburst after a well defined on time.

These prior art methods use ways to define the duration of the burst ontime and burst repetition time, but the prior art methods do not defineanything about the switching frequency within the burst on time.

Using the low power mode as described above, the operating frequency inthe low power mode can occur in the audible region for output powerlevels below a certain limit. It can be considered disadvantageous for auser to be able to hear the resonant converter operating in the lowpower mode of operation. According to an embodiment of the invention, alow power mode can be adapted in order to compare an operating frequencywith a certain audible limit (for example 18 kHz or 20 kHz), and preventthe operating frequency from dropping below that certain limit during aburst mode.

A more detailed description of this combination of features and also thecriteria for switching between low power mode and high power mode andthe corresponding settings for the converted energy per are givenhereafter.

-   -   Below a certain power level, the switching frequency gets below        the audible region, which in some examples can be considered as        20 kHz. Calculations and measurements show that a burst mode        with a burst of pulses according to the low power mode, with a        repetition frequency just above the audible region gives less        audio noise than a burst mode using pulses according to the high        power mode with the same average power level.

The combination of a burst mode that is prevented from dropping below anaudible frequency limit (such as 20 kHz) along with other featuresdisclosed herein can also be considered as an embodiment of theinvention. A specific implementation where the operating frequencyduring the burst on time of the resonant converter is limited to a lowerborder/threshold that is close to the audible region of the human ear inorder to improve audio noise performance offers significant advantagesover the prior art because it has been found that the amount of audionoise can be reduced by choosing a switching frequency during the burston time close to the upper side of the audio region. Audio noiserequirements, for example in the power adapter market, are becomingincreasingly significant.

It will be appreciated that the burst mode of operation disclosed hereinin relation to a resonant converter can also be applied with any othertype of converter that is known to operate in burst mode. A flybackconverter is one example of a converter that can benefit from theadvantages of a burst mode described herein.

In one embodiment, the functionality of FIG. 14 can be developed toinclude a burst mode of operation. The inclusion of such a burst mode ofoperation is illustrated graphically in FIG. 15.

When the resonant converter is operating in the low power mode ofoperation, and the required output power continues to decrease, thefrequency is regulated down by Vcontrol until the frequency reaches aminimum. This minimum can be set by a timer with a period time accordingto the desired frequency minimum that is started at the beginning of alow power cycle. When the period time of the low power cycle becomeslonger than the timer period, the timer triggers the start of the nextlow power cycle, preventing the frequency from becoming lower than thelimit. Assuming a fixed converted energy per cycle setting, this meansthat the minimum power cannot be reduced further. This is indicated inFIG. 15 by the horizontal line 1502 at the left side of the curve. Infact the system has now entered the burst on time. There are differentpossibilities for stopping the burst on time. This can be done by knownprior art ways such as:

-   -   burst on time determined by error loop, for example Vcontrol        crossing a minimum level;    -   fixed burst on time.

In all cases at the end of the burst on time, the converter is stoppeduntil Vcontrol causes the next low power cycle to start. There areseveral ways to implement this, and two non-limiting examples are:

-   -   (burstmode1) During the burst on time the frequency limit is        active, while during the burst off time the frequency limit is        disabled. In this case Vcontrol will drop during the burst on        time due to the surplus of power, automatically giving infinite        period time as soon as the burst on time is stopped due to the        Tper=F(Vcontrol) relation. As soon as Vcontrol rises again as        result of a dropping output voltage the infinite period time        will automatically be reduced until the next low power cycle        starts. The start of the next low power cycle triggers the next        burst on time, activating the frequency limit again.    -   (burstmode2) The frequency limit is always active, while during        the burst off time the start of the next low power cycle is        prevented, stopping the conversion. In this case Vcontrol will        drop during the burst on time, but here infinite period time        occurs as soon as the burst on time is stopped, stopping the        converter. As soon as Vcontrol rises again as result of a        dropping output voltage a Vcontrol threshold level is needed        that triggers the next burst on time when it is crossed,        switching on again the converter.

There may be provided a resonant converter including

-   -   a high power mode where the converted energy per cycle can be        adapted and    -   a low power mode where the converted energy per cycle and the        period time of a cycle can be independently adapted, and    -   means for switching between low power mode and high power mode        and setting the energy per cycle and switching frequency of the        low power mode to levels for getting optimum behaviour.

There may be provided means to make the converted energy per cycle afunction of a control variable in high power mode.

The resonant converter may further include means to make the convertedenergy per cycle, or the period time, or both, a function of a controlvariable in low power mode.

The resonant converter may further include means to let the changeoverpoint from high power mode to low power mode depend on the value of acontrol variable and a minimum time interval to stay in high power mode,which time eventually might be zero.

The changeover between high power mode and low power mode may only beallowed during a specific sub-state of the high power mode sequence. Thechangeover between high power mode and low power mode may be followed bya specific sub-state of the low power mode sequence.

The resonant converter may further include means to let the changeoverpoint from low power mode to high power mode depend on the combinationof the value of a control variable and a minimum time interval to stayin low power mode, which time eventually might be zero.

The changeover between low power mode and high power mode may only beallowed during a specific sub-state of the low power mode sequence. Thechangeover between low power mode and high power mode may always befollowed by the same specific sub-state of the high power mode sequence.

The converted energy per complete cycle in low power mode may be equalto the converted energy per complete cycle in high power mode at thechangeover point from high power mode to low power mode, scaled with apredetermined factor.

The period time of the low power mode sequence may be a function of acontrol variable. The period time of the low power mode sequence may bedependent on the actual length in time of the energy conversioninterval.

The period time of the low power mode sequence may be limited to aminimum determined by a subinterval of the low power mode (36) or aminimum determined by a clamp level at the control variable.

The control variable may depend on the error signal from the regulationloop.

Hysteresis may be present in the value of the control variable betweenthe changeover point from low power mode to high power mode and thechangeover point from high power mode to low power mode.

Within the low power mode, a burst mode may be included such that alower limit of the switching frequency of the low power cycles is set,while the burst mode is activated when the lower limit of the switchingfrequency is reached. The burst mode may operate by the sequence of:

-   -   The burst on time starts when        -   burstmode1) a timer with a time setting as function of a            control variable defining the period time of the low power            cycles during the burst off time elapses or        -   (burstmode2) The control variable crosses a level    -   At the beginning of the burst on time a burst on timer defining        the burst on time is triggered    -   The burst on time stops when the burst on timer elapses

The lower limit may be associated with the audible range of the humanear. As an example, the lower limit may be between about 15 kHz andabout 20 kHz, although it will be appreciated that any value can be usedin order to prevent the converter from operating at a frequency that canbe heard by the human ear.

Any kind of low power mode/high power mode combination where the energyper cycle (HP,LP) and period time (LP) can be set can be used.

It will be appreciated that although embodiments disclosed herein arebased on an LLC resonant converter, one or more aspects of thefunctionality disclosed herein can be used with a series resonantconverter operating in a continuous current mode (CCM). In addition,embodiments can be provided that use a half bridge plus an L+C+C inseries for lighting applications (such as compact fluorescent lighting(CFL)), where the lamp is in parallel with one of the capacitors forexample.

The invention claimed is:
 1. A controller for a resonant converter,wherein the controller is configured to operate the resonant converterin: a high power mode of operation by adjusting a first controlparameter to a vary an output power; and a low power mode of operationby adjusting a second control parameter to vary the output power; andwherein the controller is configured to set a value of the first controlparameter when changing over between the high power mode of operationand the low power mode of operation such that the output power issubstantially consistent during the changing over; wherein the firstcontrol parameter is an energy per cycle parameter; and wherein thecontroller is configured to multiply the first control parameter by afactor N when changing between the high power mode of operation and thelow power mode of operation, and wherein N corresponds to a ratio of aperiod time to a period of energy conversion pulses that will beemployed in the low power mode of operation.
 2. The controller of claim1, wherein the controller is configured to keep the first controlparameter constant for the low power mode of operation.
 3. Thecontroller of claim 1, wherein the controller is configured to operatethe resonant converter in the low power mode of operation by: adjustingonly the first control parameter to vary the output power for outputpower levels above a threshold level, and adjusting only the secondcontrol parameter for output power levels below the threshold level. 4.The controller of claim 1, wherein the controller is configured tooperate the resonant converter in the low power mode of operation by:adjusting both the first control parameter and second control parameterto vary the output power for output power levels above a thresholdlevel, and adjusting only the second control parameter for output powerlevels below the threshold level.
 5. The controller of claim 1, furthercomprising an input representative of a number of energy conversionpulses that are to be included in the period of energy conversion pulsessuch that the number of energy conversion pulses affects the period ofenergy conversion pulses.
 6. The controller of claim 1, wherein theperiod time is a minimum period time that can be used in the low powermode of operation.
 7. The controller of claim 1, wherein the controlleris configured to provide switch control signals to the resonantconverter to control the output power in accordance with the controlparameters of the high or low power mode of operation.
 8. The controllerof claim 1, wherein the controller is configured to apply a minimum timeinterval between successive power mode changes.
 9. The controller ofclaim 1, wherein the controller is configured to prevent a changeoverbetween high power mode and low power mode during at least one specificsub-state of a mode timing sequence.
 10. The controller of claim 1,configured to operate the resonant converter in a burst mode ofoperation, wherein the converter is configured to prevent a switchingfrequency within a burst on time from corresponding to an audiblefrequency range of the human ear.
 11. The controller of claim 1, whereinthe controller is configured to prevent a frequency of operation inburst mode from dropping below about 20 kHz.
 12. A method of operating aresonant converter, wherein the resonant converter is operable in: ahigh power mode of operation by adjusting a first control parameter tovary an output power; and a low power mode of operation by adjusting asecond control parameter to vary the output power; the methodcomprising: setting a value of the first control parameter when changingover between the high power mode of operation and the low power mode ofoperation such that the output power is substantially consistent duringthe changing over; and wherein the first control parameter is an energyper cycle parameter; and wherein the controller is configured tomultiply the first control parameter by a factor N when changing betweenthe high power mode of operation and the low power mode of operation,and wherein N corresponds to a ratio of a period time to a period ofenergy conversion pulses that will be employed in the low power mode ofoperation.